The flyback converter is based on the buck-boost converter.

Flyback regulators use a transformer to transfer energy from input to output. During S1 "on" time, energy builds up in the core due to increasing current in the primary winding. At this time, the polarity of the output winding is such that D1 is reverse biased. When S1 opens, the total stored energy is transferred to the secondary winding and current is delivered to the load. The turns ratio (N) of the transformer can be adjusted for optimum power transfer from input to output.

Peak switch current in a flyback regulator is equal to:

Notice that peak switch current can be reduced to a minimum
by using a very small value for N. This has two
negative consequences however; the switch voltage and
diode current become very large during switch off time.
For a given maximum switch voltage, optimum power
transfer occurs at VIN = 1/2V_{MAX}.

Both input ripple current and output ripple current are high in a flyback regulator, but this disadvantage is more than offset in many cases by the ability to achieve current or voltage gain and the inherent isolation afforded by the transformer. Output voltage is given by:

With any value of N, a duty cycle between 0 and 1 can be found which generates the required output. Flyback regulators can have an output voltage which is higher or lower than the input voltage. A disadvantage of flyback regulators is the high energy which must be stored in the transformer in the form of DC current in the windings. This requires larger cores than would be necessary with pure AC in the windings.

_{MAX}.

Flyback converters are able to regulate an
output voltage either higher or lower than the input voltage
by shuttling stored energy back and forth between the
windings of a transformer. During switch "on" time, all
energy is stored in the primary winding according to:
E = (I_{PRI})^{2}(L_{PRI})/2.
When the switch turns off, this energy
is transferred to the output winding. The current in the
secondary just after switch opening is equal to the
reciprocal of turns ratio (1/N) times the current in the primary
just prior to switch opening. Output voltage of a flyback
converter is not constrained by input voltage as in buck or
boost converters.

By varying duty cycle between 0 and 1, output voltage can theoretically be set anywhere from 0 to ∞. Practically, however, output voltage is constrained by switch breakdown voltage and the maximum output voltage is limited to:

V_{SNUB} = snubber voltage (see snubber details in this section)

V_{M} = maximum allowed switch voltage = 60V (LT1070)

This still allows the LT1070 to regulate output voltages of
hundreds or even thousands of volts by using large values
of N.

In many applications, N can vary over a wide range without
degrading performance. If maximum output power is
desired however, N can be optimized:

A second important transformer parameter which must be
determined is primary inductance (L_{PRI}). For maximum
output power, L_{PRI} should be high to minimize magnetizing
current, but this can lead to unacceptably large core
sizes. A reasonable design approach is to reduce the value
of LPRI to the point where primary magnetizing current
(ΔI) is about 20% of peak switch current. The LT1070 is
rated for 5A peak switch current, so for full power applications,
ΔI can be set to 1A peak-to-peak. Maximum output
current is reduced by one-half of the ratio of ΔI to peak
switch current, or ≅10% in this case.

With this design approach, L_{PRI} is found from:

Values of L_{PRI} higher than this will raise maximum output
current only slightly and will require larger core size.
Lower primary inductance may be used for lower output
currents to reduce core size.

I_{P} = maximum LT1070 switch current

E = overall efficiency ~ 75%

The 75% efficiency number comes from losses in the snubber network (~6%), LT1070 switch (~4%), LT1070 driver (~3%), output diode (≈8%) and transformer (~4%). Although this efficiency is not as impressive as the 85% to 95% obtainable with simple buck or boost designs, it is more than justified in many cases by the ability to use the variable N to generate high output currents or high output voltages and the option to add extra windings for multiple outputs.

The core must be able to handle **xxxx A** peak current in the
**xxxx uH** primary winding without saturating.

R1 and R2 set output voltage:

R3 and C2 provide a pole-zero frequency compensation. For details, see the section on frequency compensation elsewhere in this application note.

Flyback converters using transformers require a clamp to
protect the switch from overvoltage spikes. These spikes
are created by leakage inductance in the transformer.
Leakage inductance (L_{L}) is modeled as an inductor in
series with the primary winding which is not coupled to the
secondary as shown in Figure below.

During switch "on" time, a current is established in L_{L}
equal to peak primary current (I_{PRI}). When the switch
turns off, the energy stored in L_{L}, **(E = I ^{2}_{pri} * L_{L}/2)** will cause
the switch voltage to fly up to breakdown if the voltage is
not clamped.

If a Zener diode is used for clamping, Zener clamp voltage is selected by assigning a maximum switch voltage and maximum input voltage:

If a Zener diode is used for clamping, Zener clamp voltage
is selected by assigning a maximum switch voltage and
maximum input voltage:

The standard LT1070 maximum switch voltage is 65V, so
V_{M} is typically set at 60V to allow a margin of 5V. If we
use **V _{IN(MAX)} = xxxV **entered above, for this circuit:

Peak Zener current is equal to peak primary current (I_{PRI})
and average power dissipation is equal to:

An important part of this equation is the term [V_{Z} – (V_{OUT}
+ V_{F})/N] in the denominator. This voltage is defined as
snubber voltage (V_{SNUB}) and is the difference between the
Zener voltage and the normal flyback voltage of the primary.
(See waveforms with Figure 22.) If V_{SNUB} is too low,
Zener dissipation rises rapidly. A reasonable minimum for
VSNUB is 10V.

Leakage inductance in a transformer can be minimized by
bifilar winding or by interleaving the primary and secondary.
If this is done correctly, leakage inductance is usually
less than 1% of primary inductance. If we wind T1 for
LPRI = 230µH, L_{L} should be less than 2.3µH.

Zener dissipation under short-circuit conditions is calculated from the same equation by assuming that
VOUT = 0V and I_{PRI} is the current limit value of the LT1070.
If we let I_{PRI} = 9A: